Application Note AN-944
Use Gate Charge to Design the Gate Drive Circuit for Power
MOSFETs and IGBTs
Table of Contents
Page
1. Input behavior of a MOS-gated transistor ......................................................... 1
2. Test Circuit........................................................................................................ 1
3. The Gate Charge Curve.................................................................................... 3
4. Beware When Comparing Different Products ................................................... 4
Designers unfamiliar with MOSFET or IGBT input characteristics begin drive circuit design by
determining component values based on the gate-to-source, or input, capacitance listed on the
data sheet. RC values based on the gate-to-source capacitance normally lead to a gate drive that
is hopelessly inadequate.
Although the gate-to-source capacitance is an important value, the gate-to-drain capacitance is
actually more significant—and more difficult to deal with—because it is a non-linear capacitance
affected as a function of voltage; the gate-to-source capacitance is also affected as a voltage
function, but to a much lesser extent.
AN-944
(v.Int)
Use Gate Charge to Design the Gate Drive Circuit for Power MOSFETs and IGBTs
Topics covered:
•
•
•
•
•
Background
Test method
How to interpret the gate charge curve
How to estimate switching times
How to compare different devices
1.
Input behavior of a MOS-gated transistor
Designers unfamiliar with MOSFET or IGBT input characteristics begin drive circuit design by determining component values
based on the gate-to-source, or input, capacitance listed on the data sheet. RC value based on the gate-to-source capacitance
normally lead to a gate drive that is hopelessly inadequate.
Although the gate-to-source capacitance is an important value, the gate-to-drain capacitance is actually more significant—and
more difficult to deal with—because it is a non-linear capacitance affected as a function of voltage; the gate-to-source capacitance
is also affected as a voltage function, but to a much lesser extent. This gate-to-drain capacitance function is similar to that found
in vacuum tube amplifiers.
The gate-to-drain capacitance effect is akin to the “Miller” effect, a phenomenon by which a feedback path between the input
and output of an electronic device is provided by the interelectrode capacitance. This affects the total input admittance of the
device which results in the total dynamic input capacitance generally being greater than the sum of the static electrode
capacitances. The phenomenon of the effects of the plate impedance and voltage gain on the input admittance was first studied in
vacuum tube triode amplifier circuits by John M. Miller. Essentially, at high frequencies where the grid-to-plate (gate-to-drain)
capacitance is not negligible, the circuit is not open but involves a capacitance that is a function of the voltage gain. Solving for
the "Miller" effect is not exactly a straightforward process, even with vacuum tubes where much is known, but is even more
difficult in MOSFETs. In actuality, the gate-to-drain capacitance though smaller in static value than the gate-to-source
capacitance, goes through a voltage excursion that is often more than 20 times that of the gate-to-source capacity. Therefore, the
gate-to-drain or “Miller” capacitance typically requires more actual charge than the input capacitance.
To account for both gate-to-source and gate-to-drain capacitance in a way readily usable by designers, International Rectifier
supplies
a
“gate
charge” specifications
.68µ
µ
F
for its IGBTs and
+
HEXFET
POWER
I
D
SET
100Ω
Ω
9V
100K
MOSFETs. that can
2W
-
be used to calculate
.1µ
µ
F
drive
circuit
+20V
+20V
HEXFET 1
requirements.
Gate
C1
.1µ
µ
F
charge is defined as
22pF
.1µ
µ
F
100pF
5V
I
D
MONITOR
the charge that must
5-10 MSEC
1W5301
+V
+
500Hz
be supplied to the
8
- D1
gate, either to swing
5
1
3
1/2
7
DUT
DS0026
the gate by a given
1N414B
4.7K
1K
100Ω
Ω
V-
Ω
51Ω
amount, or to achieve
4
U1A
full switching.
U1B
I
G
V
GS
2.
Test Circuit
Figure 1.
HEXFET POWER MOSFET Gate Charge Circuit.
A typical test circuit
that can be used to measure the gate charge is shown in Figure 1. In this circuit, an approximately constant current is supplied to
the gate of the device-under-test from the 0.1 microfarad capacitor C1, through the regulator diode D1. A constant current in the
drain circuit is set by setting the voltage on the gate of HEXFET POWER MOSFET 1, so the net measurement of the charge
consumed by the gate is relative to a given current and voltage in the source-to-drain path.
AN-944
(v.Int)
OPW 2
VZR 0
2V
2µS
µ
1 WFM
2 WFM WFM
2µS
µ
OPW 0
VZR 0
2V
0 WFM
0 WFM WFM
OPW 3
VZR 0
2V
2µS
µ
An oscillogram of the gate-to-source voltage during testing,
shown in Figure 2, relates the gate voltage to time. Since a
constant current is supplied to the gate, the horizontal time
scale is directly proportional to the charge supplied to the
gate. With a suitable scaling factor, therefore, this oscillogram
is a plot of gate voltage versus charge. The point on the
oscillogram of the second voltage rise indicates where the
device is fully switched on. During the first voltage rise, the
gate-to-source capacitance is charging, and during the flat
portion, the gate-to-drain capacitance is charging. This
oscillogram therefore clearly differentiates between the charge
required for the gate-source and gate-to-drain (“Miller”)
capacitances. At the second voltage rise, both capacitances are
charged to the extent needed to switch the given voltage and
current. A more detailed explanation of the interpretation of
this data is given later. The graph in Figure 3 represents gate
voltage versus gate charge in nanocoulombs for an IRF130.
Although the second voltage rise indicates the point at which
the switching operation is completed, normal design safety
margins will dictate that the level of drive voltage applied to
the gate is greater than that which is just required to switch
the given drain current and voltage. The total charge
consumed by the gate will therefore in practice be higher than
the minimum required-but not necessarily significantly so.
For example, the gate charge required to switch 12 amps at 80
volts is 15 nanocoulombs (point A), and the corresponding
gate voltage is about 7 volts.
If the applied drive voltage has an amplitude of 10 volts (i.e.
a 3 volt margin), then the total gate charge actually consumed
would be about 20 nanocoulombs, (point B). As shown on the
graph, whether switching 10 volts or 80 volts in the drain
circuit, there is a much less than proportional difference in the
charge required. This is because the “Miller” capacitance is a
nonlinear function of voltage, and decreases with increasing
voltage.
C
V
DS
= 10V
10V
80V
80V
B
14
2 WFM
3 WFM WFM
12
10
Figure 2.
Gate Charge Waveform for Different Values of Drain
Voltage (IRF130: l
G
= 1.5 mA, I
D
=1A, V
DD
= 10, 40 and 80 volts).
The importance of the gate charge data to the designer is illustrated
as follows. Taking the charge are required to switch a previous
example, about 15 nanocoulombs of gate if 1.5 amps is supplied to the
gate, the device will be drain voltage of 80 volts and a drain current
of 12 amps. Since the 15 nC gate charge is the product of the gate
input current and the switching time, switched in 10 nS. It follows
that if 15 mA is supplied to the gate, then switching occurs in 1 us,
and so on. These simple calculations immediately tell the designer the
trade-offs between the amount of current available from the drive
circuit and the achievable switching time. With gate charge known,
the designer can develop a drive circuit appropriate to the switching
time required.
Consider a typical practical example of a 100 kHz switcher, in which
it is required to achieve a switching time of 100 nanoseconds.
V
GS
VOLTS
8
A
6
4
I
D
= 1A
2
I
D
= 12A
0
5
10
15
20
25
30
Q
G
NANOCOULOMBS
Figure 3.
Gate Voltage Versus Gate Charge for the
IRF130.
AN-944
(v.Int)
The required gate drive current is derived by simply dividing the gate charge, 15 X 10
-9
, by the required switching time, 100 X
10
-9
, giving 150 mA. From this calculation, the designer can further arrive at the drive circuit impedance. If the drive circuit
applies 14 volts to the gate, for instance, then a drive impedance of about 50 ohms would be required. Note that throughout the
“flat” part of the switching period (Figure 3), the gate voltage is constant at about 7 volts. The difference between the applied 14
volts and 7 volts is what is available to drive the required current through the drive circuit resistance.
The gate charge data also lets the designer quickly determine average gate drive power. The average gate drive power, P
DRIVE
, is
Q
G
V
G
f. Taking the above 100 kHz switcher as an example, and assuming a gate drive voltage V
G
of 14 volts, the appropriate
value of gate charge Q
G
is 27 nanocoulombs (point C on Figure 3). The average drive power is therefore 27 X 10
-9
X 14 X 10
5
=
0.038 Watts. Even though the 150 mA drive current which flows during the switching interval may appear to be relatively high.
the average power is minuscule (0.004%) in relation to the power being switched in the drain current. This is because the drive
current flows for such a short period that the average power is negligible. Thus actual drive power for MOSFETs is minute
compared to bipolar requirements, which must sustain switching current during the entire ON condition. Average drive power, of
course, increases at higher frequencies, but even at 5 MHz it would be only 1.9W.
3.
The Gate Charge Curve
The oscillograms of the gate-to-source voltage in Figure 2 neatly delineate between the charge required for the gate-to-source
capacitance, and the charge required for the gate-to-drain, or “Miller” capacitance. The accompanying simplified test circuit and
waveform diagram ( Figures 4 and 5 respectively) give the explanation. Before time t
0
, the switch S is closed; the device under
test (DUT) supports the full circuit voltage, V
DD
, and the gate voltage and drain current are zero. S is opened at time t
0
; the gate-
to-source capacitance starts to charge, and the gate-to-source voltage increases. No current flows in the drain until the gate
reaches the threshold voltage. During period T
1
to t
2
, the gate-to-source capacitance continues to charge, the gate voltage
continues to rise and the drain current rises proportionally. So long as the actual drain current is still building up towards the
available drain current, I
D
, the freewheeling rectifier stays in conduction, the voltage across it remains low, and the voltage across
the DUT continues to be virtually the full circuit voltage, V
DD
. The top end of the drain-to-gate capacitance C
AD
therefore
remains at a fixed potential, whilst the potential of the lower end moves with that of the gate. The charging current taken by C
AD
during this period is small, and for practical purposes it can be neglected, since C
AD
is numerically small by comparison with
G
CS
.
At time t
2
, the drain current reaches I
D
, and the freewheeling rectifier
+V
DD
shuts off; the potential of the drain now is no longer tied to the supply
voltage, V
DD
. The drain current now stays constant at the value I
D
enforced by the circuit, whilst the drain voltage starts to fall. Since the
gate voltage is inextricably related to the drain current by the intrinsic
transfer characteristic of the DUT (so long as operation remains in the
I
D
“active” region), the gate voltage now stays constant because the
“enforced” drain current is constant. For the time being therefore, no
further charge is consumed by the gate-to-source capacitance, because
the gate voltage remains constant. Thus the drive current now diverts,
C
DG
D
in its entirety, into the “Miller” capacitance C
AD
, and the drive circuit
charge now contributes exclusively to discharging the “Miller”
G
capacitance.
The drain voltage excursion during the period t
2
to t
3
is relatively large,
and hence the total drive charge is typically higher for the “Miller”
S
I
G
S
C
GS
capacitance C
AD
than for the gate-to-source capacitance G
CS
. At t
3
the
drain voltage falls to a value equal to I
D
x R
DS(ON)
, and the DUT now
comes out of the “active” region of operation. (In bipolar transistor
Figure 4.
Basic Gate Charge Test Circuit
terms, it has reached “saturation.”
The gate voltage is now no longer constrained by the transfer characteristic of the device to relate to the drain current, and is free
to increase. This it does, until time t
4
, when the gate voltage becomes equal to the voltage “behind” the gate circuit current
source. The time scale on the oscillogram of the gate-to-source voltage is directly proportional to the charge delivered by the
drive circuit, because charge is equal to the product of current and time, and the current remains constant throughout the whole
sequence. Thus the length of the period t
0
to t
1
represents the charge Q
GS
consumed by the gate-to-source capacitance, whilst the
length of the period t
2
to t
3
represents the charge Q
GD
consumed by the gate-to-drain or "Miller" capacitance. The total charge at
time t
3
is the charge required to switch the given voltage V
DD
and current I
D
. The additional charge consumed after time t
3
does
not represent “switching” charge; it is simply the excess charge which will be delivered by the drive circuit because the amplitude
of the applied gate drive voltage normally will be higher (as a matter of good design practice) than the bare minimum required to
accomplish switching.
AN-944
(v.Int)
4.
Beware When Comparing Different
Products
Manufacturers sometimes make technical claims for
their products that appear to be plausible, but which
in actuality do not stand up to scrutiny. A case in
point concerns the input capacitance of a power
MOSFET. Statements such as “the input capacitance
of device Y is less than that of device X, ergo Y is a
faster switch than X”, are frequently bandied about,
but are just as frequently erroneous. Apart from the
obvious speciousness of many such statements —
“apples” are frequently not compared with “apples”,
and obviously larger chips have more self capacitance
than smaller ones—the more basic fundamentals are
generally overlooked. As this application note shows,
of “bottom line” importance is the total gate charge
required for switching. The lower the charge, the
lower is the gate drive current needed to achieve a
given switching time.
A general comparison between hypothetical
MOSFETs brands “X” and “Y” is illustrated in the
Figure. Device X has a higher input capacitance;
hence the initial slope of its gate charge characteristic
is less than that of device Y. Q
GS
of device X is,
however, about the same as that of device Y, because
it has a higher transconductance and therefore requires less
voltage on its gate for the given amount of drain current
(V
GX
is less than V
GY
) The “Miller” charge consumed by
device X is considerably less than that consumed by device
Y. The overall result is that the total charge required to
switch device X, Q
X
, is considerably
less
than that required
to switch device Y, Q
Y
. Had the comparison between devices
X and Y been made on the more superficial basis of input
capacitances, it would have been concluded— erroneously—
that Y is “better” than X. Another consideration is the
energy required for switching. Again, device X scores
handsomely over device Y in this example. The energy is the
product of the gate charge and the gate voltage, and is
represented by the area of the rectangle whose corner lies at
the “switching point”. (Point 1 for device X, and point 2 for
device Y.) It is obvious that X requires significantly less gate
energy than Y. To summarize: beware of superficial
comparisons. Check the full facts before deciding which
MOSFET really has the edge in switching performance.
Q
GS
Q
GD
V
G
GATE
VOLTAGE
V
G(TH)
t
0
t
1
t
2
DRAIN
VOLTAGE
t
3
t
4
t
DRAIN CURRENT
V
DD
I
D
WAVEFORM
Figure 5.
Basic Gate Charge Waveforms
V
G
Q
Y
Q
DGY
Q
DGY
DEVICE Y
DEVICE X
V
GX
0
Q
DGX
Q
X
1
2
AREA OF THIS
RECTANGLE
(0,2 DIAGONAL CORNERS)
IS GATE ENERGY
FOR DEVICE Y
Q
AREA OF THIS
RECTANGLE
(0,1 DIAGONAL CORNERS)
IS GATE ENERGY
FOR DEVICE X
V
GY
Figure 6. Comparison of Gate Charge Characteristics of Different
Device Types.
Related Topics:
Gate drive considerations for IGBT modules
Gate drive characteristics of IGBTs
Gate drive requirements of MOS-gated transistors
High-voltage gate drive ICs
Three-phase gate drive IC
Gate drive IC for ballasts
Transformer-isolated gate driver
stm32/stm8
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