AN2123
Application Note
TD351 Advanced IGBT Driver
Principles of operation and application
by Jean-François GARNIER & Anthony BOIMOND
1 Introduction
The TD351 is an advanced IGBT driver with integrated control and protection functions. It is a simplified
version of the TD350, available in an SO8 or DIP8 package. The TD35x family (including the TD350,
TD351 and TD352) provides a wide range of drivers specially adapted to drive 1200 V IGBTs with current
ratings of 15 to 75 A in Econopak-like modules (see
Figure 2).
The main features of the TD351 are:
-
-
-
-
1 A sink/0.75 A source peak output current minimum over the full temperature range (-20°C to
125°C),
active Miller clamp function to reduce the risk of induced turn-on in high dV/dt conditions, and in
most cases, without requiring a negative gate drive,
optional 2-step turn-off sequence to reduce over-voltage in case of an over-current or a short-
circuit situation; a feature that protects the IGBT and avoids RBSOA problems,
input stage compatible with both an optocoupler and a pulse transformer.
Applications include three-phase full-bridge inverters such as in motor speed control and UPS systems
(see
Figure 1).
Figure 1. TD351 in 3-phase inverter application (1200 V IGBTs)
V+ DCbus
High-side power
supply
or
Bootstrap
Circuitry
TD351
TD351
TD351
Phase 1
Phase 2
Phase 3
Low-side power
supply
TD351
TD351
TD351
V- DCbus
AN2123/0205
Revision 1
1/15
TD351 application example
Figure 2. IGBT modules
AN2123
2 TD351 application example
A TD351 application example is shown in
Figure 3.
In this example the device is supplied by a +16V
isolated voltage source. An optocoupler is used for input signal galvanic isolation. The IGBT is driven by
44Ω for turn-on and 22Ω for turn-off thanks to the use of two gate resistors and one diode: sink and
source currents can therefore be tuned independently to help and solve EMI issues. Power switch drivers
are used in very noisy environment and decoupling of the supplies should be cared. In the application
example the decoupling is made by a 100nF ceramic capacitor located as close as possible to the TD351
in parallel with a bigger electrolytic capacitor.
Figure 3. TD351 application example
16V
TD351
1
2
4K7
16K
3
4
470pF
VH
10K
IN
VREF
CD
LVOFF
VH
OUT
VL
CLAMP
8
7
6
5
100nF
22Ω
22Ω
100pF 10nF
11V
2/15
AN2123
Input stage
3 Input stage
The TD351 is compatible with the use of both pulse transformers or optocouplers. The schematics shown
in
Figure 4
can be considered as example of use with both solutions.
When using a pulse transformer, a 2.5 V reference point can be built from the 5 V VREF pin with a
resistor bridge. The capacitor between the Vref and the bridge middle point provides decoupling of the
2.5 V reference, and also insures a high level on IN input at power-up, in order to start the TD351 in the
OFF state.
When using an optocoupler, the IN pin can be pulled-up to Vref. The pull up resistor is to be chosen
between 5 kΩ to 20 kΩ depending on the characteristics of the optocoupler. An optional filtering capacitor
can be added in case of a highly noisy environment, although the TD351 already includes filtering on
input signals and rejects signals smaller than 135 ns (t
onmin
specification).
Waveforms from the pulse transformer must comply with the t
onmin
and V
ton
/V
toff
specifications (see
Figure 5).
To turn TD351 output on, the input signal must be lower than 0.8 V for 220 ns minimum.
Conversely, the input signal must be higher that 4.2 V for 220 ns minimum in order to turn off TD351
output. A pulse width of about 500 ns at the threshold levels is recommended. In all cases, input signal at
the IN pin must be between 0 and 5 V.
Figure 4. Application schematic (pulse transformer at left; optocoupler at right)
Pulse transformer
TD351
1
10K
2
IN
VREF
Optocoupler
TD351
1
2
4K7
IN
VREF
10K
100pF 10nF
10nF
10K
Figure 5. Typical input signal waveforms with pulse transformer (left) or optocoupler (right)
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Output stage
AN2123
4 Output stage
The output stage is able to sink/source about 1.7 A / 1.3 A typical at 25°C with a voltage drop VOL/VOH
of 6 V (see
Figure 6).
The minimum sink/source currents over the full temperature range (-20°C/+125°C)
are 1 A sink and 0.75 A source. VOL and VOH voltage drops at 0.5 A are guaranteed to 2.5 V and 4 V
maximum respectively, over the temperature range (see
Figure 7).
This current capability sets the limit of
IGBT driving, and the IGBT gate resistor should not be lower than about 15Ω.
Figure 6. Typical Output stage current capability at 25°C (VH=16V)
OUT source current versus voltage (turn-on)
2.5
2.5
OUT sink current versus voltage (turn-off)
2.0
2.0
Iout (A)
1.0
Iout (A)
1.5
1.5
1.0
0.5
0.5
0.0
0
5
Vout (V)
10
15
0.0
0
5
Vout (V)
10
15
Figure 7. Typical VOL and VOH voltage variation with temperature
High level output voltage vs. Temperature
4.0
3.0
Low level output voltage vs. Temperature
3.0
VH-VOH (V)
VOL-VL (V)
Iosource=500mA
2.0
2.0
Iosink=500mA
1.0
1.0
Iosource=20mA
Iosink=20mA
0.0
-50
-25
0
25
50
Temp (°C)
75
100
125
0.0
-50
-25
0
25
50
75
100
125
Temp (°C)
4/15
AN2123
Active Miller clamp
5 Active Miller clamp
The TD351 offers an alternative solution to the problem of Miller current in IGBT switching applications.
Traditional solutions to the Miller current problem are:
l
to drive the IGBT gate to a negative voltage in OFF-state in order to increase the safety margin
l
or, to implement an additional capacitor between the IGBT gate and collector as described in the left-
hand schematic in
Figure 8)
The solution proposed by the TD351 uses a dedicated CLAMP pin to control the Miller current. When the
IGBT is off, a low impedance path is established between IGBT gate and emitter to carry the Miller
current, and the voltage spike on the IGBT gate is greatly reduced (see the right-hand schematic in
Figure 8).
The CLAMP switch is open when the input is activated and is closed when the actual gate
voltage goes close to the ground level. In this way, the CLAMP function doesn’t affect the turn-off
characteristics, but simply keeps the gate at a low level during the entire off-time.
The main benefit is that negative supply voltage can be avoided in most cases, allowing for the use of a
bootstrap technique for the high-side driver supply, and a consistent cost reduction for the application.
In addition, the use of the active Miller clamp feature avoids the need to implement any additional
capacitors between the IGBT gate and the collector. Such capacitors would negatively affect the ability of
the driver to control turn-on and turn-off.
Figure 8. Active Miller Clamp: principles of operation
High-side
driver
High-side
TD351
Miller current
Miller current
Low-side
driver
high dV/dt !
Low-side
TD351
high dV/dt !
10R
active clamp
10R
10nF
no need for
additional
capacitor
optional capacitor
implemented to
reduce voltage spike
voltage spike on IGBT gate !
reduced voltage spike
The test results shown in
Figure 9
prove how the active Miller clamp results in a consistent reduction of
the voltage spike on IGBT gate.
The left-hand waveform shows the result of a 400 V switching with a 10 nF additional Gate to Emitter
capacitor to control the voltage spike on gate.
5/15
测试/测量
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