AN1621
APPLICATION NOTE
300W SECONDARY CONTROLLED TWO-SWITCH
FORWARD CONVERTER WITH L5991A
1
INTRODUCTION
A typical off-line isolated switch-mode power supply has the controller located on the primary side of the trans-
former, whereas the output voltages to be controlled and the housekeeping functions are located on the sec-
ondary side. Usually the voltage feedback signal is transferred to the primary controller by using an optocoupler
or a transformer.
We want to propose here an asymmetrical half bridge forward converter with the controller located on the sec-
ondary side. This solution offers some advantages:
– direct use of the on-board voltage reference
– no need of the opto feedback, with its temperature and ageing gain dependence
– available on-board housekeeping functions eliminates the need of an additional dedicated device
– negligible extra cost on the gate drive transformer to satisfy safety requirements
The controller operating on the secondary side requires a specific concept for the start-up sequence, realised
here with a very simple, low consumption and cost effective solution.
2 POWER SUPPLY DESCRIPTION
2.1
Topology
Considering the 300W of output power delivered to the load, the most appropriate topology is the asymmetrical
half bridge converter.
This topology requires two power MOS transistors with a voltage breakdown equal or little higher than the max.
rectified mains voltage (thanks to the two clamping diodes), with a proper R
dson
to reach the target efficiency.
Fig. 1 shows the complete schematic diagram of the 300W Power Supply.
2.2
Start-up circuit
As already mentioned in the introduction section, the controller is located on the secondary side of the power
transformer, and for this reason, a start-up circuit has to be provided for a correct system activation.
The start-up circuit is based on a diac sending a train of controlled pulses to the low side drive section; the float-
ing drive section is energised by the second secondary gate drive transformer winding .
As soon as the L5991a wakes-up, it generates a pwm signal enabling the start-up circuit.
2.3
Gate driver
The two power mosfets, T1 and T2, are driven by a small transformer designed to satisfy the isolation safety
requirements and fast switching times.
For optimum magnetic coupling (required by the high switching frequency operation) and minimum number of
turns, a high permeability core has been selected.
AN1621/1104
Rev. 2
1/14
T1 STW14NK50
D19
1N4148
10T
50T
1T
R30 180
D13
STTH1L06
D14 STTH1L06
D15 STTH1L06
D2
IN4148
32T
R8
430
D9
18V
C16 390pF
ISEN
R15 10
Tsf2
10T
PGND
C14
470nF
C30
2.2nF
Y1
L6 9T D5 1N4148
D3
IN4148
R1 3.6
R2
10
R14
22K
C3
100pF
D16
1N4148
T5 BC327
10T
D17
1N4148
T4 BC327
T2 STW14NK50
R16
360K
DIAC
DB3
VC
13
OUT
9
VCC
8
5
FB
6
COMP
2/14
BYV52-200
10T
1T
77930
C4
1000µF
C3
1000µF
C2
1000µF
C5
1µF
P1
10K
D18
1N4148
Tsf1
C6
1nF
50T
R24
30K
D1
L1 24T
220VAC
EMI
AN1621 APPLICATION NOTE
C17 100nF
24V
13A
Vo
C26-27
220µF
400V
R12
6.2K
R11
1.2K
R9
3.3K
R10
270
C18
100nF
BRIDGE
KBUBJ
C15 100nF
R7 80K
C29
3 3nF
C7
100µF
C9 1nF
Figure 1. Schematic Diagram of 300W Power Supply
R23 180
R19
30K
DIS
14
R20
7.5K
10
11
SGND
12
SS
7
L5991A
2
RCT
C10
2.2nF
V
REF
R5 4.7K
4
DCL
15
16
ST-BY
T3
BC337
C25
100nF
IN455_MOD
C28
330pF
D12
1N4148
C12
1nF
C11
4.7µF
AN1621 APPLICATION NOTE
Table 1. Power Supply Specification
Symbol
V
I
V
O
I
O
I
l
P
o
f
s
η
Description
Input voltage
Output voltage
Output Current
Limiting Current
Output continuos power
Switching Frequency
Target Efficiency (full load)
Parameter
220Vac (176 Vac to 265 Vac) // 50Hz
24V % (ripple voltage <1%)
13Amax. continuous, 0.5Amin
constant type till ground
312W
200kHz
= 90% (from mains to output)
The core is E20/10/5-3C85, 10 turns/winding, no air-gap.
A high permeability core minimises the magnetising current to maintain a correct operation far away of the core
saturation point, at maximum duty cycle and high core temperature.
The proposed gate drive circuit allows the use of the 1:1:1 turns ratio drive transformer; moreover, a controlled
drain current rise time (by R15) and a fast fall time are achieved. A very short min. Ton pulse give the possibility
to stabilise the output voltage at max. mains and min. load.
2.4
Power MOS selection
The two-switch topology allows to use the power elements with a voltage breakdown equal to the max. rectified
mains voltage.
The mosfet used here is the STW14NK50; this device, with 500V of BVdss give us also some safety margin.
The basic parameters of the STW14NK50 are listed below:
Rdson (25°C) = 0.38Ω max., at Id = 7A (0.76Ω max. at 100°C)
Coss = 300pF typ., Qg = 75nC typ., package in TO-247
At min. supply voltage and max. load current, the conduction losses for each transistor are:
Pcon = I
2
rmsp · Rdson(100°C) = 2.23
2
· 0.76 = 3.8W where the effective Irmsp
2
is calculated in the power trans-
former section.
Estimating in about 3.2W switching and parasitic losses, the total power losses of each transistor are about 7W.
Considering 100°C of maximum operating junction temperature (at 40°C of ambient temperature) and a thermal
resistance junction-heatsink of 0.66°C/W, a heatsink of 4°C/W is required to dissipate both the transistors.
2.5
Current sense
Considering the output current rating of 13A continuous, it's our opinion that a current transformer for current
sensing is the best approach for maximising the efficiency, reliability and internal ambient temperature in case
the power supply has to be housed in a plastic box.
Due to the constant current limiting requirements, as shown in Fig2, a couple of current transformers have been
used; one transformer is sensing the current flowing into D1 (in conduction when T1 and T2 are ON) and the
second one is sensing the current flowing into D5, recirculation diode.
Oring the two transformers by D2 and D3, and closing the loop with a proper impedance value, R1, we realise
a voltage signal reproducing exactly the inductor current shape.
The current sensing loop is closed by R1 selected according the transformers turns.
Two small toroid ferrite cores (41005-TC, Magnetics, F material, 3000µ) have been used, with 50 turns.
R1 is defined by:
50
⋅
1V
R1
= ------------------
lpk
where:
1V is the nominal threshold voltage of the current sense.
Ipk is the inductor peak current( considering a 20% of current ripple, Ipk = Io +
∆I/2
= 13 + 1.3 = 14.3A)
3/14
AN1621 APPLICATION NOTE
Figure 2. Output Current Limiting Characteristic Using Two Current Transformer
Vo
25
20
15
10
5
0
D96IN456
8
10
12
14
Io
The calculated value is R1=3.5Ω
Fig. 2 shows the constant current characteristic using two current transformers.
The difference from the two current values, at output short-circuit and at current limiting intervention, is propor-
tional to the half of chocke ripple current.A choke with higher value or higher switching frequency, can reduce
this difference. If constant current feature is not requested to be constant till the output is reaching zero V, one
single current transformer can be used.
The new limiting current characteristic and the schematic diagram are shown in Fig 3a and 3b:In order to reduce
the peak current before hiccup intervention, an offset can be superimposed to the current sensing circuit to an-
ticipate the hiccup limiting current intervention, by using the additional network shown below ( figs. 4a and 4b):
Using this solution, also the
∆I/∆Vo
is higher, reducing the difference from the intervention point and the short
circuit current values.This solution can be used with two current transformers too.
Figure 3. Output Current Limiting Characteristic Using One Current Transformer
Vo
25
20
15
10
5
0
D2
R8
430Ω
R1 3.5Ω
D96IN457
1T
C6
10T
R2
D1
L1
50T
D5
C16
330pF
13
ISEN
4
6
8
10
12
14 Io
L5991A
D96IN458_mod
a:
Output current limiting characteristic
using one current transformer.
b:
limiting current schematic diagram
4/14
AN1621 APPLICATION NOTE
Figure 4. Output Current Limiting Characteristic with Foldback
1T
D1
L1
P1
Vo
25
D96IN459
50T
C6
10T
R2
D5
C4
C3
C2
3K
3K
20
R11
1.2K
15
D2
R1
120K
10
R8
5
ISEN
C16
2K
V
REF
0
13
4
4
6
8
10
12
14 Io
D96IN460_mod
L5991A
5
FB
a:
Output current limiting characteristic
b:
Schematic diagram of the modified
hiccup limiting current thereshold.
2.6
Output Diode selection
The reverse voltage of the output diodes is given by the formula:
Vr = Vin max/n = 375/3.3= 114V,
where n is the transformer turns ratio.
For a correct functionality a ultra-fast recovery diode is requested, mainly to limit switching losses and EMI prob-
lems.
To calculate the conduction losses the following equation has been applied to the selected type, BYV52-200.
For the single diode the formula is:
P = 0.7 · I
(AV)
+ 0.0075 · I
2(RMS)
that, rearranged to take into account both the diodes, becomes:
P = 0.7 · I
OMAX
+ 0.0075 · I
O2MAX
= 10.4W
where I
OMAX
= 13A.
Considering the TO247 package (1.2 °C/W total thermal resistance junction-heatsink), to ensure that the junc-
tion temperature does not exceed 100 °C at 40 °C max. ambient temperature, the heatsink has to be dimen-
sioned for about 4°C/W.
2.7
Power transformer design
The forward transformer delivers energy from the primary to the secondary without any storage.
The only consideration is with regards of the magnetising current, that has to be limited at a safety value far from
core saturation.
Our core selection is based on AP, area product, defined as AP = Aw · Ae.
5/14
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