AN-732
APPLICATION NOTE
One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106 • Tel: 781/329-4700 • Fax: 781/326-8703 •
www.analog.com
Universal Precision Op Amp Evaluation Board in SOIC Package
by Giampaolo Marino, Soufiane Bendaoud, and Steve Ranta
INTRODUCTION
The EVAL-PRAOPAMP-1R is an evaluation board which
accommodates single op amps in SOIC packages. It is
meant to provide the user with multiple choices and
extensive flexibility for different applications circuits
and configurations. This board is not intended to be
used with high frequency components or high speed
amplifiers. However, it provides the user with many
combinations for various circuit types including active
filters, differential amplifiers, and external frequency
compensation circuits. A few examples of application
circuits are given in this application note.
C7
R7
LOW-PASS FILTER
Figure 1 is a typical representation of a first-order low-
pass filter. This circuit has a 6 dB per octave roll-off
after a close-loop –3 dB point defined by f
C
. Gain below
this frequency is defined as the magnitude of R7 to R2.
The circuit might be considered as an ac integrator for
frequencies well above f
C
; however, the time domain
response is that of a single RC, rather than an integral.
f
C
= 1/(2
f
L
= 1/(2
R7
R2
C7); –3 dB frequency
C7); unity gain frequency
Acl = –(R7/R2); close loop gain
R6 should be chosen equal to the parallel combination
between R7 and R2 in order to minimize errors due to
bias currents.
R7
V
IN
R2
V
OUT
R6
V
IN
1
V
IN
2
R2
R4
R6
V
OUT
60
Figure 2. Difference Amplifier
40
f
C
20
0
f
L
–20
f
R7/R2 = 100
10f
100f
1000f
RELATIVE FREQUENCY
10000f
DIFFERENCE AMPLIFIER AND PERFORMANCE
OPTIMIZATION
Figure 2 shows an op amp configured as a difference
amplifier. The difference amplifier is the complement
of the summing amplifier, and allows the subtraction
of two voltages or the cancellation of a signal common
to both inputs. The circuit shown in Figure 2 is useful
as a computational amplifier in making a differential
to single-ended conversion or in rejecting a common-
mode signal. The output voltage V
OUT
is comprised of
two separate components:
1. A component V
OUT
1 due to V
IN
1 acting alone (V
IN
2
short-circuited to ground.)
2. A component V
OUT
2 due to V
IN
2 acting alone (V
IN
1
short-circuited to ground.)
GAIN (dB)
Figure 1. Simple Low-Pass Filter
REV. A
AN-732
The algebraic sum of these two components should be
equal to V
OUT
. By applying the principles expressed in
the output voltage V
OUT
components, and by letting R4
= R2 and R7 = R6, then:
V
OUT
1 = V
IN
1 R7/R2
V
OUT
2 = –V
IN
2 R7/R2
V
OUT
= V
OUT
1 + V
OUT
2 = ( V
IN
1 – V
IN
2) R7/R1
Difference amplifiers are commonly used in high
accuracy circuits to improve the common-mode rejec-
tion ratio, typically known as CMRR.
For this type of application, CMRR depends upon how
tightly matched resistors are used; poorly matched resis-
tors result in a low value of CMRR.
To see how this works, consider a hypothetical source
of error for resistor R7 (1 – error). Using the superposi-
tion principle and letting R4 = R2 and R7 = R6, the output
voltage would be as follows:
I
IN
1
CURRENT-TO-VOLTAGE CONVERTER
Current may be measured in two ways with an opera-
tional amplifier. Current can be converted to a voltage
with a resistor and then amplified or injected directly
into a summing node.
R7
V
OUT
R6
V
OUT
= I
IN
1
R7
Figure 3. Current-to-Voltage Converter
Figure 3 is a typical representation of a current-to-voltage
transducer. The input current is fed directly into the sum-
ming node and the amplifier output voltage changes to
exactly the same current from the summing node through
R7. The scale factor of this circuit is R7 volts per amps.
The only conversion error in this circuit is I
BIAS
, which is
summed algebraically with I
IN
1.
R7
V
OUT
R
7
R
2
+
2
R
7
error
1
−
R
2
+
R
7
×
2
R
2
=
½
R
7
VD
+
R
2
+
R
7
×
error
V
DD
=
V
IN
2
−
V
IN
1
C9
R4
V–
V+
V
OUT
From this equation, A
CM
and A
DM
can be defined as
follows:
A
CM
= R7/(R7 – R2)
A
DM
= R7/R2
error
error/2]}
L+
R9
Figure 4. Bistable Multivibrator
{1 – [(R2+2R7/R2+R7)
These equations demonstrate that when there is not an
error in the resistor values, the A
CM
= 0 and the amplifier
responds only to the differential voltage being applied to
its inputs; under these conditions, the CMRR of the circuit
becomes highly dependent on the CMRR of the amplifier
selected for this job.
As mentioned above, errors introduced by resistor
mismatch can be a big drawback of discrete differential
amplifiers, but there are different ways to optimize this
circuit configuration:
1. The differential gain is directly related to the ratio R7/
R2; therefore, one way to optimize the performance
of this circuit is to place the amplifier in a high gain
configuration. When larger values for resistors R7 and
R6 and smaller values for resistors R2 and R4 are se-
lected, the higher the gain, the higher the CMRR. For
example, when R7 = R6 = 10 k , and R2 = R4 = 1 k , and
error = 0.1%, CMRR improves to better than 80 dB. For
high gain configuration, select amplifiers with very
low I
BIAS
and very high gain (such as the AD8551,
AD8571, AD8603, and AD8605) to reduce errors.
2. Select resistors that have much tighter tolerance and
accuracy. The more closely they are matched, the better
the CMRR. For example, if a CMRR of 90 dB is needed,
then match resistors to approximately 0.02%.
BL+ = V
TH
L–
BL– = V
TL
Figure 5. Output Response
GENERATION OF SQUARE WAVEFORMS USING A
BISTABLE MULTIVIBRATOR
A square waveform can be simply generated by arrang-
ing the amplifier for a bistable multivibrator to switch
states periodically as Figure 5 shows.
Once the output of the amplifier reaches one of two pos-
sible levels, such as L+, capacitor C9 charges toward this
level through resistor R7. The voltage across C9, which
is applied to the negative input terminal of the ampli-
fier denoted as V–, then rises exponentially toward L+
with a time constant = C9R7. Meanwhile, the voltage
–2–
REV. A
AN-732
at the positive input terminal of the amplifier, denoted as
V+ = BL+. This continues until the capacitor voltage
reaches the positive threshold V
TH
, at which point
the bistable multivibrator switches to the other stable
state in which V
O
= L– and V+ = BL–. This is shown in
Figure 5.
The capacitor then begins to discharge, and its voltage,
V–, decreases exponentially toward L–. This continues
until V– reaches the negative threshold V
TL
, at which time
the bistable multivibrator switches to the positive output
state, and the cycle repeats itself.
It is important to note that the frequency of the square
wave being generated, f
O
, depends only on the external
components being used. Any variation in L+ will cause
V+ to vary in proportion, ensuring the same transition
time and the same oscillation frequency. The maximum
operating frequency is determined by the amplifier
speed, which can be increased significantly by using
faster devices.
The lowest operating frequency depends on the practical
upper limits set by R7 and C9.
Using the name convention outlined on the PRA OPAMP
evaluation board, the circuit should be connected as
follows:
B = R4/(R4 + R9); feedback factor (noninverting input)
T = 2R7
C9
ln((1 + B)/(1 – B)); period of oscillation
R
L
= 10k
C
L
= 2nF
VOLTAGE (200mV/DIV)
GND
TIME (10 s/DIV)
Figure 8. Capacitive Load Drive with Resistor
EXTERNAL COMPENSATION TECHNIQUES
Series Resistor Compensation
The use of external compensation networks may be
required to optimize certain applications. Figure 6 is a
typical representation of a series resistor compensa-
tion for stabilizing an op amp driving capacitive load. The
stabilizing effect of the series resistor isolates the op amp
output and the feedback network from the capacitive
load. The required amount of series resistance depends
on the part used, but values of 5 to 50 are usually
sufficient to prevent local resonance. The disadvantages
of this technique are a reduction in gain accuracy and
extra distortion when driving nonlinear loads.
f
O
= 1/T; oscillation frequency
R02
V
IN
C
L
R
L
V
OUT
V
IN
R
S
C
S
C
L
R
L
V
OUT
Figure 6. Series Resistor Compensation
R
L
= 10k
C
L
= 2nF
Figure 9. Snubber Network
VOLTAGE (200mV/DIV)
GND
VOLTAGE (200mV/DIV)
TIME (10 s/DIV)
GND
TIME (10 s/DIV)
Figure 7. Capacitive Load Drive Without Resistor
Figure 10. Capacitive Load Drive Without Snubber
REV. A
–3–
AN-732
R
L
= 10k
C
L
= 500pF
R
S
= 100
C
S
= 1nF
VOLTAGE (200mV/DIV)
TIME (10 s/DIV)
Figure 11. Capacitive Load Drive with the Snubber
*THESE
COMPONENTS ARE BY DEFAULT
SHORTED WITH A SMALL COPPER
STRAP BETWEEN THE MOUNTING PADS.
TO USE THE COMPONENT MOUNTING
LOCATION, SIMPLY REMOVE THE STRAP
WITH A CUTTING UTENSIL (DREMEL
CUTTING TIP, EXACTO BLADE, ETC.)
C2
*R1
V1
R
T
1
G1
0
R5
R3
C5
2
C7
R7
VCC
*R11
C1
R8
*C8
ALL PASSIVE COMPONENT MOUNTING
LOCATIONS CAN ACCOMMODATE 0805 TO 2512
SIZED SURFACE MOUNT PACKAGES, 1/2" SPACED
LEADED RESISTORS AND 0.3" SPACED
LEADED CAPACITORS
C9
R10
C10
G3
*R2
DUT
7
V+
6
*R
O
*R
O
1
*R
O
2
VO
R
S
C
S
C
L
R
L
G5
3
V–
4
R9
V2
R
T
2
G2
0
*R12
*R4
R6
C6
C11
G4
VEE
Figure 12. EVAL-PRAOPAMP-1R Electrical Schematic
Figure 13. EVAL-PRAOPAMP-1R Board Layout Patterns
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners.
–4–
REV. A
AN04894–0–8/04(A)
Snubber Network
Another way to stabilize an op amp driving a capacitive
load is with the use of a snubber, as shown in Figure 9.
This method presents the significant advantage of not
reducing the output swing because there is not any
isolation resistor in the signal path. Also, the use of
the snubber does not degrade the gain accuracy or
cause extra distortion when driving a nonlinear load.
The exact R
S
and C
S
combinations can be determined
experimentally.
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