®
AN1453
APPLICATION NOTE
NEW FAMILY OF 150V POWER SCHOTTKY
By F. GAUTIER
INTRODUCTION
Nowadays, the Switch Mode Power Supply
(SMPS) is becoming more widespread as a result
of computer, telecom and consumer applications.
The constant increase in services (more
peripherals) and performance, which offers us
these applications, tends to move conversion
systems towards higher output power.
In addition to these developments dictated by the
market, SMPS manufacturers are in competition,
their battlefield being the criteria of power density,
efficiency, reliability and cost, this last being factor
very critical.
Today, SMPS designers of 12V-24V output have
practically the choice between a 100V Schottky or
a 200V bipolar diode.
The availability of an intermediate voltage has
become necessary to gain in design optimization.
This is why STMicroelectronics is introducing a
new family of 150V POWER SCHOTTKY diodes,
intended for 12V and more secondary rectification,
in applications such as desktops, file servers or
adaptors for notebook.
Consequently, this application note will underline
the advantages of a 150V Schottky technology
compared to a 200V ultra fast diode.
In order to do this, the example of a Flyback
converter will be used, and the static and dynamic
parameters of the 150V Schottky will be detailed,
as well as their influence in this converter.
1. CONDUCTION LOSSES & EFFICIENCY GAIN
Schottky diodes are mainly used for output
rectification. In a typical SMPS working with a
switching frequency lower than 100kHz,
conduction losses are generally the main losses in
the diode. They are directly linked to the curve of
forward voltage (V
F
) versus forward current (I
F
),
and obviously the best gain in efficiency will be
obtained with the lowest V
F
.
July 2001
In the following examples, the conduction losses
between a 150V Schottky and a 200V bipolar
diode in a Flyback and a Forward converter will be
compared.
The conduction losses in the diode are calculated
from the classical formula:
2
P
cond
= V
T0
⋅
I
F(AV)
+ R
d
⋅
I
IF(RMS)
V
t0
:threshold voltage with V
F(@ IF)
= V
T0
+ R
d
.I
F
R
d
: dynamic resistance with R
d
=
∆
V
F
/
∆
I
F
where V
T0
and R
d
are calculated from the current
range of current view by the diode (Fig. 1), for
better accuracy.
Figure 1 shows also, the typical current through
the rectification diode and the corresponding I
F(AV)
and I
2
IF(RMS)
:
Fig. 1:
Typical current through a rectification diode
I
D
I
ma x
I
min
t
0
αI
D
.T
T
α
ID
(I
max
+
I
min
)
2
α
ID 2
2
I
2
(I
max
+
I
min
+
I
max
⋅
I
min
)
F(RMS)
=
3
V
−
V
F(@ Imin)
R
d
=
F(@ Imax)
V
T0
=
V
F(@ imax)
−
R
d
⋅
I
max
I
max
−
I
min
I
F(AV)
=
NB:
-In the datasheet, the V
T0
and R
d
are maximum
values given for I
F
and 2 I
F
at 125°C.
-In discontinuous mode I
min
=0.
1/9
APPLICATION NOTE
1.1. Example 1: FLYBACK
The first example is a 24V/48W Flyback converter
working in continuous mode (Vmains=90V) with
the following conditions:
α
ID
=
0.4, I
max
ID
=
6.66A, I
min
ID
=
3.33A, I
out
=
2A
Fig. 2:
Rectification diode in a Flyback converter
1.2. Example 2: FORWARD
In the following example, the conduction losses in
a 12V/96W Forward converter are simulated:
Fig. 4:
Rectification diode in a Forward converter
D1
V
in
D2
I
L
I
o u
I
D
V
in
I
o u t
α
D1
=
0.3, I
Lmax
=
9A, I
Lmin
=
7A, I
out
=
8A
Calculations per diode give:
I
F(AV)per diode
= 1A and I
F(RMS) per diode
= 1.6A
We can now calculate the efficiency gain (∆η(%)=
η
ref
-
η)
for this Flyback converter which has a
reference (ref) efficiency of 85% with
STPR1020CT:
Fig. 3:
Example of efficiency gain in Flyback
converter
P
out
=48W
V
out
=24V
V
T0
R
d
typ(V) mΩ
1.5A, 3A,
125°C
0.58 46.5
∆P
(W)
η=85
%
∆η%
Calculations per diode give:
I
F(AV)D1
=
2.4A, I
F(RMS)D1
=
4.39A
I
F(AV)D2
=
5.6A, I
F(RMS)D2
=
6.71A
The difference of efficiency between a
STPR1620CT (2x8A, 200V Ultrafast) and a
STPS16150CT (2x8A, 150V Schottky) for a 12V
output, are given in table Fig. 5:
Fig. 5:
Example of efficiency gain in Flyback
converter
V
T0
R
d
typ(V) mΩ
7A, 9A,
125°C
0.8
20
20
η=85
%
∆η%
Ref
P
cond
(W)
P
out
=96W
V
out
=12V
P
cond
(W)
∆P
(W)
STPR102CT
2x5A / 200V
PN diode
STPR162CT
2x8A / 200V
PN diode
1.4
0 (ref) 0 (ref)
STPR1620CT
6.48
5.60
Ref
STPS16150CT 0.68
0.54 46.5
1.32
-0.08 +0.12
-0.95 +0.72
STPS10150CT
2x5A / 150V
0.50
Schottky diode
STPS16150CT
2x8A / 150V
0.47
Schottky diode
43
1.22
-0.18 +0.27
These two examples show that whatever the type
of converter, a significant efficiency gain can be
achieved only by replacing a 200V bipolar diode by
a 150V Schottky.
40
1.14
-0.26 +0.39
2/9
APPLICATION NOTE
2. REVERSE LOSSES AND T
JMAX
2.1. Reverse losses: Prev
The reverse losses can be determined by:
P
rev
=
V
R
⋅
I
R
⋅
(1
− α
)
with:
(1- ):
duty cycle when the reverse voltage (V
R
) is
applied
I
R
:
leakage current versus V
R
and operating
junction temperature (T
j
)
V
R
:
reapplied voltage accross the diode
Fig. 6 shows an example of reverse losses in a
Flyback converter with the following conditions:
(1−
α
)
=
0.4, V
R
=
80V, T
j
=
125
°
C
Fig. 6:
Example of reverse losses in a Flyback
converter
I
Rtyp
per diode
100V, 125°C
STPS10150CT
130µA
P
rev
per diode
4.2mW
Example:
Flyback converter with 2 diodes in parallel
(1
− α
)
=
0.4, c
=
0.069, V
R
=
80V
R
th(j
−
c)total
=
2.4
°
C / W, R
th(c
−
a)
=
7.6
°
C / W
Fig. 7:
Example of T
jmax
with STPS10150CT
For a dual
diode
STPS10150CT
I
R(VR,Tjmax)
I
Rmax
(80V, 125°C)
1.3mA
T
jmax
45.28mA 176.5°C
This example shows that in a typical application, a
150V Schottky can be used up to 175°C.
STMicroelectronics specifies in the datasheet
T
jmax
at 175°C.
3. SWITCHING BEHAVIOUR
3.1. Turn-on behaviour
The behaviour at turn-on is characterized by a low
value of peak forward voltage (V
FP
) and forward
reverse recovery time (t
fr
) (Fig. 8).
Fig. 8:
V
FP
and t
fr
for STPS16150CT
I
F
=16A
dI
F
/dt=100A/µs
T
j
=25°C
Per diode
STPS16150CT
t
fr
(ns)
V
FP
(V)
Thus, the reverse losses are very low due to the
low value of the leakage current.
The following paragraph will show that due to
these low values of reverse current, the thermal
runaway limit is only reached for high junction
temperature.
2.2. T
jmax
before thermal instability is reached
Remembering that the stability criterion is given
by:
dP
rev
1
<
dT
j
R
th(j
−
a)
with:
P
rev
=
V
R
.I
R(VR,Tjmax)
.(1
− α
)
The above formulae give the critical value of the
leakage current before the thermal runaway limit is
reached:
1
I
R(VR,Tjmax)
=
V
R
⋅
c.R
th(j
−
a)
⋅
(1
− α
)
The evolution of the leakage current versus T
j
and
V
R
is given by:
I
R(V
R
,Tj)
=
I
R(V
R
,125)
exp
c(Tj
−
125)
From these physical laws, it can be deduced that:
T
jmax
=
125
+
I
R(V
R
,Tjmax)
1
⋅
In
c
I
R
max
(V
R
,125
°
C)datasheet
100
2.2
These values depends mainly on the dI
F
/dt. The
switching losses at turn-on are always negligible.
3.2. Turn-off behaviour
The turn-off behaviour is a transitory phenomenon
(ns), but repetitive depending on the switching
frequency. It is a source of spike voltage, noise
and for high switching frequency, of non-negligible
switching losses.
In order to illustrate this phenomenon, the example
of a Flyback converter will be used once again.
The difference in behaviour between a 150V
Schottky and 200V bipolar diode will be compared
for the three following points: spike voltage, EMC
and switching losses.
3/9
APPLICATION NOTE
3.2.1. Difference of spike voltage between a
150V Schottky and 200V PN diode
In a Flyback converter, the reverse voltage (V
R
used in §2) across the diode will be maximum, for
the maximum mains voltage (V
INmax
):
V
R
=
V
INmax
⋅
n
s
+
V
out
n
p
(cf Fig. 9)
3.2.1.1 Turn-off behaviour for a PN diode
In the datasheet are specified the main turn-off
parameters (Q
rr
, I
RM
, t
rr
…). These parameters are
represented in Fig. 10:
Fig. 10:
Key parameters at turn-off for a bipolar
diode without snubber
Q
rr
= Q
rra
+ Q
rrb
t
rr
= t
a
+ t
b
t
S=
b
t
a
I
In addition to this nominal reverse voltage (V
R
),
generally an overvoltage spike at the turn-off of the
diode is observed (Fig. 9). It can be shown that
with a conventional bipolar diode, this spike is
more important for a Flyback converter working in
continuous mode than in a discontinuous mode.
In the case of a high spike voltage, the Maximum
Repetive Reverse Voltage (V
RRM
) of the diode has
to be oversized, compared with the real need (V
R
)
defined in Fig. 9.
To limit this peak and to preserve a "guard band"
with the V
RRM
(in order to avoid reaching the
breakdown voltage), the designer places a
snubber circuit (R
S
, C
S
) in parallel with the diode.
Generally, the "guard band" is such that the
maximum voltage reapplied to the diode does not
exceed 80% of the V
RRM
.
Fig. 9:
Spike voltage across the rectification diode
d
I F
/dt
d
I R
/dt
V
Q
rra
I
RM
t
a
t
b
t
Q
rrb
V
R
V
Rma x
The following oscillogram shows the turn-off
behaviour for a bipolar diode (STPR1620CT) with
snubber and without snubber, in a 24V/45W
Flyback working in continuous mode.
To observe the phenomenon correctly, it is
necessary to compensate the delay time between
the voltage and the current, (by temporal shift) due
to the measuring equipment Fig. 11.
Fig. 11:
Switching behaviour of a 200V bipolar
diode
dI
F
/dt=130A/µs
R
s
C
s
n
p
V
in
n
s
I
D
V
s
V
P
V
D
I
D
V
o u t
V
D
V
R
= V
IN
n
s
+ V
out
n
p
V
Rmax
Turn-off diode
I
RM
t0
Tj=100°C
dI
R
/dt=600A/µs
I
V
I
V
RRM
I
RM
=4A
delay time
Compensative curve
V
V
Rmax
=250V
R
s
=22ohms
C
s
=2.2nF
I
This spike voltage is due to the leakage inductance
of the transformer (L
f
) and to the nature of the
recovery charge of the diode, which itself depends
on the diode technology: bipolar diode or Schottky
diode.
t0
dV/dt
I
V
R
=42V
V
V
20V/div
2A/div
50ns/div
V
Rmax
=90V
4/9
APPLICATION NOTE
Without a snubber, in this example the diode is
repeatedly in conduction because the oscillation is
very strong. Furthermore, the voltage is close to
the breakdown voltage. This means that the
system is no longer reliable and a snubber circuit is
necessary.
On these 2 oscillograms, we can see that the value
of the maximum reverse current (I
RM
) is defined
when the reverse voltage rises (typical behaviour
of a bipolar diode). At this time the voltage is not
fixed by the diode.
The curve Q
rr
, I
RR
versus dI
F
/dt and T
j
is given in
the datasheet. For example in Fig. 12, the
evolution of I
RM
versus dI
F
/dt for a STPR1620CT
can be observed.
Fig. 12:
Peak reverse recovery current versus
dI
F
/dt (per diode)
STPR1620CG/CT
IRM(A)
20
IF=IF(av) 90% confidence
Tj=125°C
Fig. 13:
Equivalent model at t
0
for a bipolar diode
R
s
C
s
n
p
L
P
n
s
L
s
V
D
V
s
V
R
= V
S
+ V
o u t
L
f
V
o u t
I
L
=I
RM
f
C
s
V
D
C
Qrrb
C
j
R
s
10
n
s
⋅
V
IN
n
p
L
f
:
leakage inductance of the transformer
C
j
:
junction capacitance
C
Qrrb
:
equivalent capacitance modeling the
reverse charge, necessary for the establishment of
the potential barrier, which supports the reverse
voltage.
V
out
:
output voltage
V
S
=
With the following initial conditions at t=t
0
:
I
L
f
=
I
RM
bipolar
and V
D
≈
0
The equivalent schematic can be used to define
V
D
=
V
R
max
NB:
1)
W i th ou t s n u b b e r, t h e r e i s a L
f
, C c ir cu it
( C = C
j
+ C
Qr rb
) which lead to a second order
differential equation:
d
2
V
C
+ ω
2
⋅
V
C
+ ω
2
⋅
V
R
=
0 and
ω
2
=
1 / L
f
⋅
C
0
0
0
2
dt
with initial conditions at t=t
0
:
I
L
f
=
I
RM
and V
C
0
=
V
D
=
0
In this equation, an approximation is made with C
constant, because in reality C
j
and C
Qrrb
vary with
the voltage applied.
The solution of the differential equation gives us:
V
R
max
=
V
D
=
V
R
+
V
R
2
Where:
V
s
:
secondary voltage
1
10
20
50
100
dIF/dt(A/µs)
200
500
It can be also noticed, that the parameter I
RM
significantly increases with the temperature.
In continuous mode the dI
F
/dt (few hundred A/µs)
is fixed by the leakage inductance and the reverse
voltage (V
R
):
dI
F
V
R
n
with V
R
=
s
⋅
V
IN
+
V
out
=
dt
L
f
n
p
It is many time higher than in discontinuous mode
(lower than 1A/µs):
dI
F
V
out
=
withL
S
〉〉
L
f
dt
L
S
+
L
f
(L
S
: Secondary inductance)
Thus, with this curve we can see that, in
continuous mode (high dI
F
/dt), the bipolar diode
must evacuate a non-negligible charge, which
means a higher I
RM
. This is verified on oscillogram
Fig. 11.
With this value of I
RM
, an equivalent model at t
0
with a snubber circuit can be established:
L
f
+
I
RM
⋅
C
2
Therefore we can see that the V
Rmax
depends the
leakage inductance (L
f
) and on recovery charge
(I
RM
). Thus, V
R
max
is very dependent on the
temperature.
5/9
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