AN1214
APPLICATION NOTE
DESIGN TIPS FOR L6561 POWER FACTOR CORRECTOR
IN WIDE RANGE
by Cliff Ortmeyer & Claudio Adragna
This application note will describe some basic steps to optimize the design of the L6561 PFC for wide
range voltage input (105V- 300V) while also having a broad output power range (65W - 105W). Initial
design steps are covered in application note AN966. This is to serve as a supplement to that applica-
tion note and also give an example of a wide range demo board optimized for the US market (110V -
277V). A deeper look at the control of the L6561 can also be found in application note AN1089 “Control
loop modeling of L6561-based TM PFC”.
Introduction
Designing a PFC circuit with a singular input voltage and a singular output power is a task that is rather straight
forward and gives a very good set of component values when the design equations are used. The task becomes
a little more difficult when a wide range PFC is needed and the specifications are tight. This is common in ap-
plications such as lighting where there is a demand for good power factor ( >.99) and THD less than 10% in the
full range of nominal operating conditions. The problem occurs since the design must be done for the worst
case conditions which are a low input voltage and maximum output power. As we will see, this will diminish the
performance of the PFC circuit when the input voltage is high and the output power is low. What must be done
in this case is to look closely at the limits of the L6561 and external components and to optimize or compromise
where needed.
Design Tips
Multiplier Operation.
Once the initial design is done and measurements have been made, the next step is to look
at the operating parameters of the L6561 to see that it is working within its full capabilities without going over its
linear operating range. A copy of the table we will be referring to is shown in Fig. 1.
Figure 1. Multiplier Characteristics
V
CS
(pin4)
(V)
upper voltage
clamp
D97IN555A
V
COMP
(pin2)
(V)
3.5
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
0
5.0
4.5
4.0
3.2
3.0
2.8
2.6
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
V
MULT
(pin3) (V)
For optimal operation the device should stay in the
linear operation of the multiplier. As can be seen,
there are three pins that should be measured in the
worst case conditions. The first is with the lowest in-
put voltage (low line) and the highest output power.
The second is at the highest input voltage (high line)
and the lowest output power. The first pin to be
measured is the Vcomp (pin2). This is the output of
the error amplifier (Figure 2) and will determine
which curve will be referenced when measuring the
other two parameters - Vcs and Vmult. Once this is
established, the peak voltage of the multiplier input
(pin 3) should be measured and noted. Next mea-
sure the peak voltage of the current sense resistor
(Vcs - pin 4). Looking at the graph in Figure 1, de-
termine which curve to use from the Vcomp voltage.
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December 2000
AN1214 APPLICATION NOTE
Next, note where the Vcs and Vmult are on the curve to make sure that they are in the linear operating region.
If operation in the linear region is not met, adjust the variable that allows linear operation to be met. If however
the device is operating in the linear region but is not allowing the full range of the multiplier to be used, then
increasing one of the variables (the multiplier voltage for example) can help to maximize the full operating range
of the multiplier.
Figure 2. Multiplier Block Diagram.
Rs
CURR.CMP
E/A
X
1.7V
Zero crossing dead time.
Once the multiplier operating parameters have been met, the input voltage as well as
the input current should be looked at together. One problem to look for is a distortion of the current waveform
especially at high line and low load. An example can be seen in figure 3.
Figure 3. Current Shape at Zero Crossing with High Capacitance FET and slow Turn-on Diode.
The main reason for this effect is that near zero-crossings the energy stored in the inductor is very low, not
enough to charge up the drain node total capacitance (basically, the FET’s drain-to-source capacitance C
oss
and the inductor’s parasitic intrawinding capacitance) to turn the boost diode on. The turn-on speed of the boost
diode adds to the problem as well. As a result, energy is exchanged between reactive components and there is
no input-to-output transfer. This can be seen in figure 3.
To minimize Coss, the Rds
(ON)
of the FET should be maximized within the limits of acceptable conduction loss-
es, and its voltage rating should be the minimum that still provides adequate breakdown capability. In fact, both
2/6
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D97IN675
-
2
3
4
AN1214 APPLICATION NOTE
a low Rds
(ON)
and a high voltage imply a higher C
oss
.
Inductor parasitic capacitance can be reduced by minimizing the number of winding layers. Adding a layer of
tape between winding layers can reduce the capacitance considerably. The use of a slotted bobbin is also very
effective.
Also optimizing the diode can offer a positive contribution. A minimum junction capacitance will be somewhat
beneficial, even though this is a minor contribution to the total drain capacitance.
A major improvement can be offered by a diode with a well controlled die resistivity (such as Turboswitch series)
which has a lower peak forward voltage, so that it actually turns on just a few volts above the PFC output voltage.
An example of the improvement given by optimizing the FET and the diode is shown in figure 4.
Figure 4. Current Shape at Zero Crossing with Lower Capacitance FET and Turboswitch Diode
Input capacitance (EMI filtering).
Another source of error can be due to the input filter. Since the voltage output
from the rectifier bridge is used as the reference for the current to follow, any distortions in this waveform will
translate into distortions of the current waveform, hence lower power factor or greater THD. One cause of this
can be due to too large of a high frequency filter capacitor being used after the bridge. A high value capacitor
can filter the rectified voltage and cause the voltage to deviate from a rectified sinusoid and even not reach zero
at light load. This can be seen in figure 5.
Figure 5. Non-discontinuous Voltage Error
The obvious way of improving this is to lower the high frequency capacitor value. Care must be taken not to
lower the capacitance such that the effectiveness of the EMI filter (in front of the diode bridge) is not degraded
so as to not pass regulatory requirements. So, by lowering that capacitance, the HF filter capacitor in front of
the diode bridge may need to be increased.
Switching frequency.
One other method of using the full dynamic range of the L6561 is to reduce the minimum
switching frequency of the FET. By using the lowest possible switching frequency of the L6561, a wider range
3/6
AN1214 APPLICATION NOTE
of switching frequencies are available to be used. This helps minimize the effects of the internal propagation
delay as well as the offset of the current sense comparator. In this way the current will track the voltage wave-
form better, in particular near the zero-crossings. This, however, must be weighed against the size increase of
the inductor because a lower switching frequency implies a larger inductance value.
When determining the lowest frequency, it must be noted that switching below 15kHz is not recommended since
this may interfere with the internal starter.
A special construction technique of the inductor can offer the optimum compromise: one that allows the use of
a low inductance value so as to minimize inductor size and, at the same time, have a lower switching frequency
near zero-crossings. The price to pay for that is an additional step in the inductor manufacturing flow.
It is the so-called “step-gap core” technique: the centre leg of one half of the ferrite core is ground so that the
air gap thickness has a step change, as shown in fig. 6.
At low inductor current the small thinner part of the air gap dictates a high inductance value (L1). As current
increases above a certain value (I
L1
), the thinner part of the air gap will progressively saturate and the induc-
tance will drop to a value (L2<L1) determined by the thicker part of the air gap (l
g
). A non-linearity is deliberately
introduced so that, for a given switching frequency at the top of the rectified sinusoid - where current is high -
(that is, for a given L=L2), the switching frequency near the zero-crossings - where current is low - will not go
as high as with a linear inductor.
Figure 6. Step-gap ferrite grinding and its effect on inductance value
b
"step-gap"
h
l
g
L
L1
core half
L2
I
L1
I
L2
I
L
The appropriate height h and breadth b of the ferrite step (both of them determine L1 and I
L1
) for a given appli-
cation will be found empirically.
4/6
AN1214 APPLICATION NOTE
Wide Range Example Circuit
An example circuit was designed with a varying voltage input of 110V to 277V and an output power of 65W to
105W. The techniques in this paper (except the step-gap core) were used to help bring the power factor and
THD into acceptable levels. The example schematic and associated EMI filter are shown below along with the
measured results.
Figure 7. Example Schematic
D1 STTA106
C6
T
R7 (*)
998K
C3 1µF
68K
5
2
1
7
4
R12
1M
+
Vo=450V
Po=105W
R11
1M
C5
47µF
250V
R3 (*)
240K
BRIDGE
+ 4 x 1N4007
FUSE 4A/250V
-
Vac
(105V to 305V)
NTC
R10
10K
D3 1N4150
D2
1N5248B
R2
100
12nF
R1
C1
33nF
630V
R9 (*)
1.24M
8
R5
10
L6561
3
C2
22µF
25V
C7
10nF
6
MOS
STP6NB50
C5'
47µF
250V
R6 (*)
0.5
1W
R8
5.6K
1%
D99IN1098
-
(*) R3 = 2 x 120KΩ
R6 = 1Ω/2
R7 = 2 x 499KΩ, 1%
R9 = 2 x 620KΩ
TRANSFORMER
T: core THOMSON-CSF B1ET2910A (ETD 29 x 16 x 10mm) OR EQUIVALENT (OREGA 473201A8)
primary 90T of Litz wire 10 x 0.2mm
secondary 7T of #27 AWG (0.15mm)
gap 1.25mm for a total primary inductance of 0.8mH
Figure 8. Emi Filter
T1
LINE
C
x
T2
PFC
C
y
EARTH
D97IN680
Table 1. Example Schematic Results
Vin
[V
AC
)
105
120
277
300
105
120
277
300
Vout
[V]
449
449
449
449
449
449
449
449
Iout
[mA]
136
136
136
136
237
237
237
237
PF
0.998
0.998
0.995
0.993
0.999
0.999
0.997
0.994
THD
[%]
4.7
5.3
9.7
11.0
4.0
4.0
7.7
11.2
Pin
[W]
66.4
65.8
65.0
65.0
115.8
114.7
111.0
111.0
Pout
[W]
61.0
61.0
61.0
61.0
106.5
106.5
106.5
106.5
Efficiency
[%]
91.9
92.7
93.9
93.9
92.0
92.9
96.0
96.0
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