SL486
APPLICATION NOTES - REFER TO FIG. 4
Diode Anode and Cathode (Pins 1 and 16)
The infra-red
receiving diode is connected between pins 1 and 16. The
input circuit is configured so as to reject signals common to
both pins. This improves the stability of the device, and greatly
reduces the sensitivity to radiated electrical noise, The diode
is reverse biased by a nominal 0·65V
Gyrator C2 and C1 (Pins 2 and 3)
The decoupling, provided
by gyrator C2 and C1, rolls off the gain of the feedback loop
which balances the DC component of the infra-red diode
current. The values of C2 and C1 are chosen to produce a low
frequency cut-off characteristic below a nominal 2kHz. Hence,
the gyrator produces approximately 20dB rejection at 100Hz.
The gyrator consists of two feedback loops operating in
tandem. Only one feedback path is functional when the DC
component of the diode current is less than 200µA. This loop
is decoupled by gyrator C2. For diode currents between
200µA and 1·5mA the second control loop is operative, and
this is decoupled by gyrator C1.
The decoupling capacitors, gyrator C2 and C1, must be
connected between pins 2 and 3, to pin 4. The series imped-
ance of C2 and C1 should be kept to a minimum.
First Stage Decouple (Pin 15)
The capacitor on pin 15
decouples the signal from the non-inverting input of the first
difference amplifier (see also Fig. 2). The capacitance of 15nF
is chosen to produce a 2kHz low frequency roll-off. The
capacitor must be connected between pins 15 and 14 (the
input ground).
Second Stage Decouple (Pin 5)
The capacitor on pin 5
decouples the signal from the non-inverting input of the
second difference amplifier. The capacifance of 33nF is
chosen to produce a 2kHz low frequency roll-off. The capaci-
tor must be connected between pins 5 and 4 (the input V
CC
).
Fourth Stage Decouple (Pin 6)
The capacitor on pin 6
decouples the signal from the non-inverting input of the fourth
difference amplifier. The capacitance of 4.7nF is chosen to
produce a 2kHz low frequency roll-off. The capacitor must be
connected between pins 6 and 7 (the output V
CC
).
AGC Decouple/Delay Adjust (Pin 8)
The output of the fourth
difference amplifier is followed by a peak detector, which is
used to provide an AGC control level. This produces a current
source which is limited to 10mA at pin 8. The AGC decoupling
capacitor (C5 normally 150nF) filters the pulsed input, and the
resultant level controls the gain of the first three difference
amplifiers.
The AGC control level exhibits a fast attack/slow decay
characteristic. Immediately infra-red pulses are detected, the
gain will be reduced, so that any weaker noise pulses that are
also received will not be seen at the output. Thus, provided the
infra-red pulses are the most intense, it is possible to receive
data in noisy environments. The slow decay keeps the AGC
level intact during data reception, and produces a delay
before any received noise may become present at the output,
when transmission ceases.
Output (Pin 9)
The output will be low, pulsing high with a
source impedance of a nominal 55kΩ , for a received infra-
red pulse. It is a linear amplification of the input and swings
between output ground and output V
CC
.
Stretch Input and Stretch Output (Pins 10 and 11)
A typical
infra-red PPM system transmits very narrow pulses. The
duration of these pulses is typically 15µs, so in order to use a
microprocessor-based decoder system it is necessary to
lengthen the received pulse. This stretched output can be
obtained from pin 11 when a capacitor is connected between
pins 9 and 10 (C8 in Fig. 4).
The width of the pulse is determined by the value of this
coupling capacitor and is defined in the Electrical Character-
istics. The stretch output is normally high, pulsing low for a
received infra-red pulse and swings between V
CCO
and V
EEO
.
It must be noted that the stretch output logic sense is
inverse to that of the output on pin 9 so must be re-inverted for
microprocessor applications.
Regulator Input, V
REGIN
(Pin 12)
The device can be operated
with supplies of between 4·5V and 9·0V connected between
input/output ground (pins 14 and 13) and input and output V
CC
(pins 4 and 7) as shown in Fig. 3. The device can also be
operated with supplies in excess of 9·0V by using the on-chip
regulator. In this case connections are made between V
CCO
(pin 7) and the regulator input V
REGIN
(pin 12) as shown in Fig.
4. A supply voltage of between 9·0V and 18V will then cause
V
EEO
(pin 13) to be regulated at a level nominally 6·4V below
V
CCO
(pin 7). The regulator will, however, lose control with a
potential difference of less than 9·0V. Below this level the
voltage on pin 13 will track nominally 1·5V above the level of
pin 12. When the regulator is not used (low voltage operation),
pin 12 must be connected to V
EEO
(pin 13).
OPERATING NOTES - REFER TO FIGS. 3 AND 4
Gyrator C1 (Pin 3)
If the environment in which the device is
operating limits the background light such that the DC compo-
nent of the diode current has a maximum of 200µA, it may be
desirable to omit (as in Fig. 3) the more bulky and costly 68µF
capacitor (gyrator C1 shown in Fig. 4). In this case pin 3 can
be left open circuit. The resultant application will then have a
characteristic of greatly reduced gain when the ambient light
causes the DC current to rise above this threshold.
Alternatively,the 68µF capacitor can be replaced by a
resistor.
The outcome of this is to further reduce the gain in ambient
light levels above the 200µA threshold. Below this threshold
the overall gain is slightly enhanced as the light level ap-
proaches the threshold value. If chosen, this resistance
should lie between 10kΩ and 200kΩ .
Noise Immunity
The stretch output can also be used as a
means of improving performance relating to a receiver sys-
tem, over and above its main purpose of providing a micro-
processor interface. Including C8 (Fig. 4) causes the output
pulses (from pin 9) to be subjected to the stretch input
threshold. Thus any noise pulses from pin 9 that are below this
threshold will not be seen at the stretch output (pin 11). A
further improvement can be made, using this stretch input
threshold, by including some additional filtering of the output
(C10 in Fig. 4). This can be adjusted in value (typically 100pF)
to reduce some of the noise pulses that otherwise cross the
threshold, to a level below the threshold.
Screening
Use of screening for the device, and associated
components, improves the performance and immunity to
externally radiated noise. The screening method used must
protect the sensitive front-end of the device; provided that
the diode, pin 1-pin 16, C2 (pin 2) and the first stage decoupling
(pin 15) are screened, it may be found that for the application
considered, the remalning circuitry need not be so protected.
In applications where externally radiated noise is minimal, it
may be possible to reduce any screening to pins 1 and 16 and
the diode connections only. Screening may not be necessary
in some instances, but this largely depends on the level of
radiated noise, the decoupling/filtering employed and the
receiver’s decoding technique.
Decoupling
Typical decoupling arrangements for use with or
without the regulator are given in Figs. 4 and 3, respectively.
When using the regulator, further improvements in high
frequency supply rejection are possible by the inclusion of R2.
The value can be chosen so as to keep the pin 12 end of R2
within the
29·0
to
218V
(wrt pin 7) specified voltage range.
For example, if the SL486 is used in a system with a supply
of 16V, a typical value tor R2 would be 200Ω. Note that the
regulator is a low impedance point between pins 12 and 13.
C7 thus maintains a low impedance path between pins 4 and
12 at high frequencies.
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